NTB52N10 ,Power MOSFET 52 Amps, 100 Volts3R DRAIN–TO–SOURCE RESISTANCE (NORMALIZED) R , DRAIN–TO–SOURCE RESISTANCE () I , DRAIN CURRENT (AM ..
NTB52N10G , N-Channel Enhancement−Mode D2PAK
NTB5404N ,Power MOSFET, 40 V, 167 A, Single N-ChannelELECTRICAL CHARACTERISTICS (T = 25°C unless otherwise stated)JParameter Symbol Test Condition Min T ..
NTB5405N ,Power MOSFET, 40 V, 116 A, Single N-ChannelELECTRICAL CHARACTERISTICS (T = 25°C unless otherwise stated)JParameter Symbol Test Condition Min T ..
NTB5405NT4G , Power MOSFET 40 V, 116 A, Single N−Channel, D2PAK
NTB5426N ,DatasheetELECTRICAL CHARACTERISTICS (T = 25°C Unless otherwise specified) JCharacteristics Symbol Test Condi ..
OPA842IDBVT ,Wideband, Low Distortion, Unity Gain Stable, Voltage Feedback Operational Amplifier.(1)ABSOLUTE
OPA842IDBVTG4 ,Wideband, Low Distortion, Unity Gain Stable, Voltage Feedback Operational Amplifier 5-SOT-23 -40 to 85OPA842
OPA842IDG4 ,Wideband, Low Distortion, Unity Gain Stable, Voltage Feedback Operational Amplifier 8-SOIC -40 to 85ELECTRICAL CHARACTERISTICS: V = ±5V (continued)SBoldface limits are tested at +25°C. At T = +25°C, ..
OPA842IDR ,Wideband, Low Distortion, Unity Gain Stable, Voltage Feedback Operational AmplifierMAXIMUM RATINGSOver operating free-air temperature range, unless otherwise noted.OPA842 UNITPower S ..
OPA842IDR ,Wideband, Low Distortion, Unity Gain Stable, Voltage Feedback Operational AmplifierFEATURES DESCRIPTION2• UNITY-GAIN BANDWIDTH: 400MHzThe OPA842 provides a level of speed and dynamic ..
OPA842IDRG4 ,Wideband, Low Distortion, Unity Gain Stable, Voltage Feedback Operational Amplifier 8-SOIC -40 to 85 SBOS267D –NOVEMBER 2002–REVISED SEPTEMBER 2010Wideband, Low Distortion, Unity-Gain Stable, Voltage ..
NTB52N10
Power MOSFET 52 Amps, 100 Volts
3R DRAIN–TO–SOURCE RESISTANCE (NORMALIZED) R , DRAIN–TO–SOURCE RESISTANCE () I , DRAIN CURRENT (AMPS)DS(on), DS(on) DR , DRAIN–TO–SOURCE RESISTANCE ()DS(on)I , LEAKAGE (nA)DSS I , DRAIN CURRENT (AMPS)DNTB52N10POWER MOSFET SWITCHINGSwitching behavior is most easily modeled and predicted The capacitance (C ) is read from the capacitance curve atissby recognizing that the power MOSFET is charge a voltage corresponding to the off–state condition whencontrolled. The lengths of various switching intervals (Δt) calculating t and is read at a voltage corresponding to thed(on)are determined by how fast the FET input capacitance can on–state when calculating t .d(off)be charged by current from the generator. At high switching speeds, parasitic circuit elementscomplicate the analysis. The inductance of the MOSFETThe published capacitance data is difficult to use forsource lead, inside the package and in the circuit wiringcalculating rise and fall because drain–gate capacitancewhich is common to both the drain and gate current paths,varies greatly with applied voltage. Accordingly, gateproduces a voltage at the source which reduces the gate drivecharge data is used. In most cases, a satisfactory estimate ofcurrent. The voltage is determined by Ldi/dt, but since di/dtaverage input current (I ) can be made from aG(AV)is a function of drain current, the mathematical solution isrudimentary analysis of the drive circuit so thatcomplex. The MOSFET output capacitance alsot = Q/IG(AV)complicates the mathematics. And finally, MOSFETs havefinite internal gate resistance which effectively adds to theDuring the rise and fall time interval when switching aresistance of the driving source, but the internal resistanceresistive load, V remains virtually constant at a levelGSknown as the plateau voltage, V . Therefore, rise and fall is difficult to measure and, consequently, is not specified.SGPtimes may be approximated by the following: The resistive switching time variation versus gateresistance (Figure 9) shows how typical switchingt = Q x R /(V – V )r 2 G GG GSPperformance is affected by the parasitic circuit elements. Ift = Q x R /Vf 2 G GSPthe parasitics were not present, the slope of the curves wouldmaintain a value of unity regardless of the switching speed.whereThe circuit used to obtain the data is constructed to minimizeV = the gate drive voltage, which varies from zero to VGG GGcommon inductance in the drain and gate circuit loops andR = the gate drive resistanceGis believed readily achievable with board mountedand Q and V are read from the gate charge curve.2 GSP components. Most power electronic loads are inductive; thedata in the figure is taken with a resistive load, whichDuring the turn–on and turn–off delay times, gate current isapproximates an optimally snubbed inductive load. Powernot constant. The simplest calculation uses appropriateMOSFETs may be safely operated into an inductive load;values from the capacitance curves in a standard equation forhowever, snubbing reduces switching losses.voltage change in an RC network. The equations are:t = R C In [V /(V – V )]d(on) G iss GG GG GSPt = R C In (V /V )d(off) G iss GG GSP6000V = 0 V V = 0 VDS GST = 25°CJ5000Ciss40003000CrssCiss20001000CossCrss0–10 –5 0 5 10 15 20 25V VGS DSGATE–TO–SOURCE OR DRAIN–TO–SOURCEVOLTAGE (VOLTS)Figure 7. Capacitance Variation
